The invention relates to the field of modulation of signals of communication systems. More particularly, the present invention relates to subcarrier modulation methods for efficient frequency reuse within exiting bandwidths.
Due to the limited availability of spectrum allocations for communications systems, it has become desirable to reuse existing spectrum by employing bandwidth efficient modulation methods to existing satellite communications systems. In particular, it has become desirable to add new signals to existing quadrature multiplexed spread-spectrum communication signals, such as the signals employed in the Global Positioning System (GPS), within the existing spectral allocation. As the number of users of a given communication system increases, it is often desirable to augment the system with additional communication signals. One method that has been used to achieve this goal in timing, telemetry and command links is conventional subcarrier phase modulation. Furthermore, it is often necessary to modulate new signals onto existing quadrature multiplexed communication systems within the existing spectral allocation. It is desirable that any such approach satisfy various constraints, including causing minimal distortion of existing signals, transmitting new signals through the same high power amplifier used by existing signals, accommodating new data messages and new pseudo random noise (PRN) code families for spread-spectrum systems, providing flexibility to control the spectral separation of signals within the allocated band, and providing flexibility to control the distribution of energy in and outside of the allocated band. Unfortunately, for the existing quadrature-multiplexed communication systems having data modulations on both the I and Q channels, that is, the in-phase channel with zero degree phase offset from the carrier, and the quadrature channel with ninety degree phase offset from the carrier, adding another signal slightly offset in frequency gives rise to a non-constant envelope which causes distortion to the existing signals when the total waveform is passed through a non-linear amplifier. For non-quadrature-multiplexed communication systems, the subcarrier modulation method has been employed to permit the realization of a constant-envelope modulation. Unfortunately, a general approach for applying the subcarrier modulation method to quadrature-modulated communication systems has not been previously developed.
GPS is undergoing a transformation with the Block IIF satellite. This redefinition of GPS from a military service with the guarantee of civil use to a true dual service is one of the GPS modernization goals. The transformation started out as a modest upgrade that involved a new civil L5 frequency and a military acquisition signal at the L2 frequency but has evolved into a complement of new signals at L1 and L2 frequencies for enhanced military and civil use. It is desirable to design and choose optimum M-code military signals and signal modulation methods that achieve better than current performances without degrading the existing signals. Under consideration are two classes of signals, the Manchester code signals and the binary offset carrier signals. These signals result from modulation of a non-return to zero pseudo random noise spreading code by a square-wave subcarrier. The Manchester code signal is a special case where the bit rate of the spreading code and the frequency of the square-wave are the same. It is also equivalent to a Manchester encoded PRN code. The binary offset carrier signal has been defined as encompassing all the other cases where the rate of the spreading code is less than the subcarrier frequency. These signals share the characteristic of conventional subcarrier modulation that the waveform exhibits a null at the carrier frequency due to the square-wave subcarrier, and therefore, can allow for the transmission of the new GPS military M-codes along with the course acquisition C/A-codes, and precision P(Y) codes, where (Y) denotes the encrypted form of the P-code. One of the key challenges for a new signal scheme for the GPS transmitter is to transmit the C/A, P(Y), and M-code signals using a flexible efficient, constant envelope modulation method, compatible with goals of the GPS modernization.
To this end, it is desirable to develop a general high-efficiency constant-envelope subcarrier modulation approach to enable quadrature multiplexing of an M-code signal or signals onto the L1 and L2 GPS carriers that is applicable to both a combined-aperture for transmitting C/A, P(Y), and M-code signals through the same upconverter and amplifier chain and antenna, and a separate-aperture for transmitting M-code signals out of a separate upconverter, amplifier chain and antenna from C/A and P(Y) signals without resulting in significant signal loss due to inefficient signal modulation methods. The current Block II and Block IIR satellites employ constant-envelope quadrature multiplexing of two bi-phase modulated ranging code signals at the GPS L1 frequency, namely the C/A code and P(Y) code. The baseline Block IIF satellite will also transmit these signals on both the GPS L1 and L2 frequencies, along with the new military signals.
The new M-code signal consists of the product of a military data modulation multiplied by a spreading code modulation. Previously proposed approaches for augmenting the existing GPS waveform, with a new split-spectrum M-code signal, such as the tri-code hexaphase modulation method, involve the linear addition of a single new M-code signal on to the existing GPS waveform. These approaches multiplex the additional signal with an existing I/Q quadrature modulation by adding a third signal to one of the I and Q phases. Unfortunately, this gives rise to the undesirable non-constant envelope, that is, a variable amplitude. For this case, the envelope of the composite signal is no longer constant due to the presence of the time-varying amplitude component in addition to the constant component of the envelope. This result is undesirable because the amplitude variations will give rise to AM-to-AM and AM-to-PM distortions when the signal is passed through a nonlinear amplifier unless the operating point of the amplifier is backed off from its saturation point to the linear region of amplifier operation. Such a back off can result in appreciable power losses.
During the course of the GPS military signal development study, a number of M-code signal modulation methods have been proposed. Two of the leading approaches were hard-limiting the sum of the C/A, P and M-code signals, and majority voting. The hard-limiting approach involves forcing the non-constant envelope waveform resulting from the sum of C/A and M on one carrier phase and P on the other, to be constant by limiting the amplitude variation to a minimum value. This technique results in significant signal power loss and distortion for the case of equal C/A and M-code power levels as is the case for hard-limited tri-code hexaphase modulation method. Furthermore, the exact efficiency is critically related to the power balance between C/A, P(Y) and M-code signals and the desired balance between signals is not easy to achieve. An alternative approach to combining M-code with C/A and P(Y) code signals is through majority vote combining. In the majority vote approach, signals are time multiplexed, that is, time-shared, on either I or Q phases to allow multiple signals to be transmitted in a single constant envelope. The disadvantage of this approach results from the relatively large majority combining loss per code in combination, assuming equal power levels for all codes in the majority combination. Furthermore, it is difficult to control the relative power levels of the combined signals without incurring additional combining losses.
Conventional subcarrier modulation has been recognized as a means to modulate additional signals onto a modulated carrier signal while maintaining a constant envelope. The Space-Ground-Link Subsystem for example, employs three subcarriers that are phase-modulated onto a carrier signal to enable the modulation of four signals, including one carrier modulation and three subcarrier modulations. The classical subcarrier approach, applied to non-quadrature-multiplexed communication systems, involves the phase-modulation of a carrier signal by one or more periodic subcarriers that are typically sine-wave, square-wave or triangle-wave signals. In this classical subcarrier approach, the range of the phase modulation is determined by a so-called modulation index, usually controlled by the gain of the phase modulator. In conventional subcarrier systems, cross-product intermodulation components of the waveform are interpreted as signal losses. A variation of conventional subcarrier modulation applicable to augmenting the existing GPS signals with new M-code signals is desirable due to the inefficiency of the previously proposed M-code signal combining techniques, that is, signal modulation methods.
An alternative combining method has been proposed to add the new M-code signals by spatially combining the new M-code signal with the C/A and P(Y) code signals by transmitting the new M-code signals through a separate antenna and amplifier chain. While this approach would, in principal, allow for the transmission of C/A-codes and P(Y)-codes within the M-code, the simplification in modulator design is more than offset by the impact on the satellite when adding an independent antenna and amplifier chain. Due to potential implementation difficulties with the separate-aperture approach, and a requirement to efficiently transmit the new M-code signals with world-wide coverage, the more challenging problem of transmitting these signals is the development of a general modulation approach to enable the transmission of these signals through a single modulator, upconverter, power amplifier chain and antenna aperture.
The leading modulation methods, that have been proposed thus far to achieve this end, have a maximum efficiency limited to roughly seventy-one percent in the allocated GPS spectral band. Efficiency is defined as the sum of the effective transmitter power after reception plus any band limiting losses and other losses associated with this sum divided by the total transmitted power. At the same time, the leading approaches with the highest efficiency restrict options available to optimize the waveform by transmitting a single M-code signal used for both acquisition and tracking.
These single M-code approaches include tri-code hexaphase, hard-limited tri-code hexaphase, majority vote code multiplexing, and offset carrier and binary offset carrier modulations. The tri-code hexaphase, offset carrier and binary offset carrier modulation approaches have a non-constant envelope. As a result, these approaches cause distortion to existing signals when the total waveform is passed through a non-linear amplifier. The majority vote modulation approach employs a constant envelope but does not provide significant spectral separation and has limited inherent flexibility in adjusting the amplitude of generated harmonics. The majority vote modulation approach also suffers from majority combining losses that results in relatively poor overall power efficiency. The same is true for the hard-limited tri-code hexaphase modulation method.
The transmission of both data and spreading code modulations in a quadrature-multiplexed communication channel is well known. Conventional quadrature phase shift keying (QPSK) direct-sequence spread spectrum modulation schemes generally employ spreading sequences on the in-phase and quadrature channels. The modulation of unspread data is a special case. For GPS L1 or L2 carriers modulated by codes and data, a carrier signal is modulated by codes and data in a conventional quadrature modulated communication signal as S0(t) in terms of the in-phase I0(t) and quadrature Q0(t) signals. The signal S0(t) can express GPS L1 and L2 carriers modulated by C/A and P codes and data in a conventional quadrature modulation communication method in terms of an S0(t)Q0/I0 equation with respect to an I0 equation and a Q0 equation.
S0(T)=I0(t)cos(xcfx89Ct)xe2x88x92Q0(T)Sin(xcfx89Ct)
                    I        0            ⁢              (        t        )              =                                        2            ⁢                          P              I                                            P            T                              ⁢                        D          I                ⁢                  (          t          )                    ⁢                        C          I                ⁢                  (          t          )                                        Q        0            ⁢              (        t        )              =                                        2            ⁢                          P              Q                                            P            T                              ⁢                        D          Q                ⁢                  (          t          )                    ⁢                        C          Q                ⁢                  (          t          )                    
The CI(t) and DI(t) terms are the in-phase pseudo random noise spreading code and data modulation, and CQ(t) and DQ(t) are the quadrature pseudo random noise spreading code and data modulations. PI and PQ are the average power of the I and Q signals in relation to total power PT.
The signal S0(t) may also be expressed in an S0(t)A0/xcfx86 equation. An A0(t) magnitude equation describes the envelope form where A0(t) is the magnitude envelope of S0(t). A xcfx86(t) equation describes the phase of S0(t).
S0(t)=A0(t)Cos(wCt+xcfx86(t))
                    A        0            ⁢              (        t        )              =                                                      I              0              2                        ⁢                          (              t              )                                +                                    Q              0              2                        ⁢                          (              t              )                                          =                                                  2              ⁢                              (                                                      P                    I                                    +                                      P                    Q                                                  )                                                    P              T                                      =        CONSTANT                        φ      ⁢              (        t        )              =                            Tan                      -            1                          ⁢                                            Q              0                        ⁢                          (              t              )                                                          I              0                        ⁢                          (              t              )                                          =                        Tan                      -            1                          ⁢                                                            P                Q                                            P                I                                              ·                                                                      D                  Q                                ⁢                                  (                  t                  )                                            ⁢                                                C                  Q                                ⁢                                  (                  t                  )                                                                                                      D                  I                                ⁢                                  (                  t                  )                                            ⁢                                                C                  I                                ⁢                                  (                  t                  )                                                                        
The total average power may be described by a PTF equation for the signal S0(t) that can be expressed by the magnitude of the envelope squared divided by two, in reference to identity equations based on the square of the CI and CQ code signals and the DI and DQ data signals.       P    TF    =                    A        0        2            2        =                                        I            0            2                    +                      Q            0            2                          2            =                                    P            I                    +                      P            Q                                    P          T                    xe2x80x83CI(t)2=CQ(t)2=DI(t)2=DQ(t)2=1
Conventional approaches to multiplexing additional signals with an existing I/Q quadrature modulation involve adding a third signal to one of the two I or Q phases. Unfortunately, this gives rise to the undesirable non-constant envelope characteristic. A signal can be linearly added to the I phase of the baseband waveform modulated onto the same carrier as the existing I/Q signals. For this case, the magnitude envelope of the signal A0(t) is no longer constant due to presence of the time-varying component of A(t) in addition to the constant component of A0(t). This result is undesirable because the amplitude variations will give rise to amplitude modulation to amplitude modulation (AM-to-AM), and amplitude modulation to phase modulation (AM-to-PH) distortions when the signal is passed through a nonlinear amplifier. These and other disadvantages are solved or reduced using the invention.
An object of the invention is to provide a constant envelope composite signal with added subcarrier signals that do not distort existing signals when the composite signal is passed through a high-power amplifier operating near saturation.
Another object of the invention is to provide for the modulation of new signals onto a quadrature multiplexed communication channel while controlling the power balance between signals while maintaining a constant envelope.
Yet another object of the invention is control the spectral separation of quadrature multiplexed signals within the allocated band through the use of subcarrier frequencies, subcarrier codes rate, and a subcarrier modulation index.
Yet another object of the invention is to enable the modulation of an orthogonal pair of subcarrier signals onto the I and Q phases of a quadrature modulated carrier.
Another object of the invention is to transmit the new M-code signals using either separate or combined-apertures with reduced distortion and losses.
A further object of the invention is to provide control of the distribution of energy among signals inside and outside an allocated band.
Yet a further object of the invention is to utilize cross-product intermodulation signals as useful communication signals.
In a general aspect of the invention, quadrature product subcarrier modulation (QPSM) enables the transmission of a quadrature multiplexed carrier modulation with one or more subcarrier signals in the same constant-envelope waveform. The generalized QPSM enables the application of subcarrier modulation to quadrature multiplexed communication systems such as quadrature phase shift keying (QPSK) or minimum shift keying (MSK). The QPSM can be applied to both direct and spread-spectrum quadrature multiplexed communication systems. In particular, QPSM is advantageous for any spread spectrum system desiring additional spread signals with spectral isolation between new and existing pseudo random noise (PRN) code signals using the same transmitter power amplifier. QPSM can augment existing two-code spread-spectrum systems, without the need to employ time multiplexing or majority voting that result in significant power losses, while maintaining a constant envelope signal with spectral separation between existing signals and new signals with high efficiency. The QPSM enables a quadrature- multiplexed subcarrier spread-spectrum waveform modulation using, in general, multiple rate product codes that cause minimal interference to existing codes and the new codes. The modulation index can be used to control the distribution of energy between carrier and subcarrier signals.
CASM is applicable to the transmission of quadrature duplexed military acquisition (MA) and military tracking (MT) signals in a flexible and efficient manner. The high efficiency approach is applied to both combined-aperture, with C/A, P(Y), MA and MT passed through the same upconverter amplifier chain and antenna and separate-aperture, with MA and MT signals transmitted out of a separate upconverter, amplifier chain and antenna from C/A and P(Y)) communication path. The CASM waveform is a highly efficient means of quadrature multiplexing new GPS military acquisition and tracking signals with flexibility in adjusting the relative power of combined signals without greatly altering the power efficiency of the GPS waveform. The combined-aperture CASM offer power-efficient constant-envelope GPS modernization waveforms including the combination of C/A, P and M-code signals. The generated M-code signals are mathematically equivalent to the spatially combined signals.
CASM employs a subcarrier to phase modulated new M-code signals on the same carrier as the current C/A and P(Y) ranging codes. The constant-envelope CASM modulation is an evolution of the constant-envelope subcarrier modulation used on SGLS and other terrestrial and space systems to quadrature-multiplexed systems. Unlike prior subcarrier modulation developments, the CASM utilizes cross-product intermodulation terms as new ranging communication signals. In the present invention, using subcarrier modulation, cross product terms are interpreted as signals and not as losses. Preferably, the M-Code signals are generated by employing square-wave subcarriers to modulate the new military ranging signals, due to ease of hardware implementation, but CASM may be generated using both squarewave and sinewave subcarriers for GPS modernization. These and other advantages will become more apparent from the following detailed description of the preferred embodiment.